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PDF OPA642 Data sheet ( Hoja de datos )
| Número de pieza | OPA642 | |
| Descripción | Wideband / Low Distortion / Low Gain OPERATIONAL AMPLIFIER | |
| Fabricantes | Burr-Brown | |
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®
OPA642
OOPPAA66452 8
OPA642
Wideband, Low Distortion, Low Gain
OPERATIONAL AMPLIFIER
FEATURES
q LOW DISTORTION: –95dBc at 5MHz
q GAIN OF +1 BANDWIDTH: 400MHz
q AVAILABLE IN SOT23-5 PACKAGE
q HIGH OPEN LOOP GAIN: 95dB
q HIGH COMMON-MODE REJECTION: 90dB
q FAST 12-BIT SETTLING: 13ns (0.01%)
q LOW NOISE: 2.7nV/√Hz
q HIGH OUTPUT CURRENT: ±60mA
q VERY LOW DIFF GAIN/PHASE ERROR:
0.007%/0.008°
APPLICATIONS
q ADC/DAC BUFFER AMPLIFIER
q LOW DISTORTION IF AMPLIFIER
q HIGH RESOLUTION IMAGING
q MEDICAL IMAGING
q LOW NOISE PREAMPLIFIER
q HIGH CMR DIFFERENCING AMPLIFIER
q TEST INSTRUMENTATION
q PROFESSIONAL AUDIO
DESCRIPTION
The OPA642 provides a level of speed and dynamic
range previously unattainable in a monolithic op amp.
Using a unity gain stable voltage feedback architec-
ture with two internal gain stages, the OPA642 achieves
exceptionally low harmonic distortion over a wide
frequency range. The “classic” differential input pro-
vides all the familiar benefits of precision op amps,
such as bias current cancellation and very low invert-
ing current noise compared with wideband current
feedback op amps. Fast settling time, excellent differ-
ential gain/phase performance, low voltage noise and
high output current drive make the OPA642 ideal for
most high dynamic range applications.
Unity gain stability makes the OPA642 particularly
suitable for low gain differential amplifiers,
transimpedance amplifiers, gain of +2 video line driv-
ers, wideband integrators and low distortion ADC buff-
ers. Where higher gain or even lower harmonic distor-
tion is required, consider the OPA643—a higher gain-
bandwidth and lower noise version of the OPA642
+5V
1Vp-p
5MHz
+5V
OPA642
2Vp-p 0.1µF
5kΩ
REFT
0.1µF
clk
ADS804
24Ω
10MSPS
Analog
Input
12-Bit
10MSPS
100pF
Measured
80dB SFDR
–5V
402Ω
402Ω
5kΩ
REFB
0.1µF
High Dynamic Range 10MSPS Digitizer
International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111
Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132
©1993 Burr-Brown Corporation
PDS-11190F
OPA642Printed in U.S.A. February, 1998
®
1 page  
TYPICAL PERFORMANCE CURVES (CONT)
At TA = +25°C, VS = ±5V, RL = 100Ω, RF = 402Ω, G = +2, unless otherwise noted. RF = 25Ω for a gain of +1.
5MHz 2ND HARMONIC DISTORTION
–65
G = +2
–70
RL = 100Ω
–75
–80 RL = 200Ω
–85
RL = 500Ω
–90
–95
–100
0.1
1
Output Voltage Swing (Vp-p)
10
5MHz 3RD HARMONIC DISTORTION
–65
G = +2
–70
–75
–80
RL = 100Ω, 200Ω, or 500Ω
–85
–90
–95
–100
0.1
1
Output Voltage Swing (Vp-p)
10
10MHz 2ND HARMONIC DISTORTION
–60
G = +2
–65
RL = 100Ω
–70
–75 RL = 200Ω
–80
–85 RL = 500Ω
–90
–95
–100
0.1
1
Output Voltage Swing (Vp-p)
10
10MHz 3RD HARMONIC DISTORTION
–60
G = +2
–65
–70
–75
–80
–85
–90
–95
–100
0.1
RL = 500Ω
RL = 200Ω
RL = 100Ω
1
Output Voltage Swing (Vp-p)
10
20MHz 2ND HARMONIC DISTORTION
–50
G = +2
–55
–60
RL = 200Ω
–65
RL = 100Ω
–70
–75
–80
RL = 500Ω
–85
–90
0.1
1
Output Voltage Swing (Vp-p)
10
20MHz 3RD HARMONIC DISTORTION
–50
G = +2
–55
–60
–65
–70
–75
–80
–85
–90
0.1
RL = 200Ω
RL = 100Ω
RL = 500Ω
1
Output Voltage Swing (Vp-p)
10
®
5 OPA642
5 Page 
In the inverting configuration, an additional design consid-
eration must be noted. RG becomes the input resistor and
therefore the load impedance to the driving source. If imped-
ance matching is desired, RG may be set equal to the
required termination value. However, at low inverting gains
the resultant feedback resistor value can present a significant
load to the amplifier output. For example, an inverting gain
of 2 with a 50Ω input matching resistor (= RG) would require
a 100Ω feedback resistor, which would contribute to output
loading in parallel with the external load. In such a case, it
would be preferable to increase both the RF and RG values,
and then achieve the input matching impedance with a third
resistor to ground. The total input impedance becomes the
parallel combination of RG and the additional shunt resistor.
BANDWIDTH VS GAIN
Voltage feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90°, as it
does in high gain configurations. At low signal gains, most
amplifiers will exhibit a more complex response with lower
phase margin. The OPA642 is optimized to give a maxi-
mally flat second order Butterworth response in a gain of 2.
In this configuration, the OPA642 has approximately 60° of
phase margin and will show a typical –3dB bandwidth of
150MHz. When the phase margin is 60°, the closed-loop
bandwidth is approximately √2 greater than the value pre-
dicted by dividing GBP by the noise gain. Increasing the
gain will cause the phase margin to approach 90° and the
bandwidth to more closely approach the predicted value of
(GBP/NG). At a gain of +10, the 21MHz bandwidth shown
in the Typical Specifications agrees with that predicted
using the simple formula and the typical GBP of 210MHz.
OUTPUT DRIVE CAPABILITY
The OPA642 has been optimized to drive the demanding
load of a doubly terminated transmission line. When a 50Ω
line is driven, a series 50Ω into the cable and a terminating
50Ω load at the end of the cable are used. Under these
conditions, the cable’s impedance will appear resistive over
a wide frequency range, and the total effective load on the
OPA642 is 100Ω in parallel with the resistance of the
feedback network. The Specifications show a guaranteed
±2.5V swing into a such a load—which will then be reduced
to a ±1.25V swing at the termination resistor. The guaran-
teed ±35mA output drive over temperature provides ad-
equate current drive margin for this load. Higher voltage
swings (and lower distortion) are achievable when driving
higher impedance loads.
A single video load typically appears as a 150Ω load (using
standard 75Ω cables) to the driving amplifier. The OPA642
provides adequate voltage and current drive to support up to
3 parallel video loads (50Ω total load) for an NTSC signal.
With only one load, the OPA642 achieves an exceptionally
low 0.007%/0.008° dG/dP error.
DRIVING CAPACITIVE LOADS
One of the most demanding, and yet very common, load
conditions for an op amp is capacitive loading. A high speed,
high open-loop gain, amplifier like the OPA642 can be very
susceptible to decreased stability and closed-loop response
peaking when a capacitive load is placed directly on the
output pin. In simple terms, the capacitive load reacts with
the open-loop output resistance of the amplifier to introduce
an additional pole into the loop and thereby decrease the
phase margin. This issue has become a popular topic of
application notes and articles, and several external solutions
to this problem have been suggested. When the primary
considerations are frequency response flatness, pulse re-
sponse fidelity and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts
it and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.
The Typical Performance Curves show the recommended
RS vs Capacitive Load and the resulting frequency response
at the load. The criterion for setting the recommended
resistor is maximum bandwidth, flat frequency response at
the load. Since there is now a passive low pass filter between
the output pin and the load capacitance, the response at the
output pin itself is typically somewhat peaked, and becomes
flat after the rolloff action of the RC network. This is not a
concern in most applications, but can cause clipping if the
desired signal swing at the load is very close to the amplifier’s
swing limit. Such clipping would be most likely to occur in
pulse response applications where the frequency peaking is
manifested as an overshoot in the step response.
Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA642. Long PC board
traces, unmatched cables, and connections to multiple de-
vices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA642 output pin
(see Board Layout Guidelines).
DISTORTION PERFORMANCE
The OPA642 is capable of delivering an exceptionally low
distortion signal at high frequencies and low gains. The
distortion plots in the Typical Performance Curves show the
typical distortion under a wide variety of conditions. Most of
these plots are limited to 100dB dynamic range. The
OPA642’s distortion does not rise above –100dBc until
either the signal level exceeds 0.5V and/or the fundamental
frequency exceeds 500kHz. Distortion in the audio band is
≤ –120dBc.
Generally, until the fundamental signal reaches very high
frequencies or powers, the second harmonic will dominate the
distortion with negligible third harmonic component. Focus-
ing then on the second harmonic, increasing the load imped-
ance improves distortion directly. Remember that the total
load includes the feedback network—in the non-inverting
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11 OPA642
11 Page | ||
| Páginas | Total 15 Páginas | |
| PDF Descargar | [ Datasheet OPA642.PDF ] | |
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