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PDF MAX1639 Data sheet ( Hoja de datos )
| Número de pieza | MAX1639 | |
| Descripción | High-Speed Step-Down Controller with Synchronous Rectification for CPU Power | |
| Fabricantes | Maxim Integrated | |
| Logotipo |  |
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19-1337; Rev 0; 2/98
EVAALVUAAILTAIOBNLEKIT
High-Speed Step-Down Controller with
Synchronous Rectification for CPU Power
________________General Description
The MAX1639 is an ultra-high-performance, step-down
DC-DC controller for CPU power in high-end computer
systems. Designed for demanding applications in which
output voltage precision and good transient response are
critical for proper operation, it delivers over 35A from 1.1V
to 4.5V with ±1% total accuracy from a +5V ±10% supply.
Excellent dynamic response corrects output transients
caused by the latest dynamically clocked CPUs. This
controller achieves over 90% efficiency by using synchro-
nous rectification. Flying-capacitor bootstrap circuitry
drives inexpensive, external N-channel MOSFETs.
The switching frequency is pin-selectable for 300kHz,
600kHz, or 1MHz. High switching frequencies allow the
use of a small surface-mount inductor and decrease out-
put filter capacitor requirements, reducing board area
and system cost.
Output overvoltage protection is enforced by a crowbar
circuit that turns on the low-side MOSFET with 100%
duty factor when the output is 200mV above the normal
regulation point. Other features include internal digital
soft-start, a power-good output, and a 3.5V ±1% refer-
ence output. The MAX1639 is available in a 16-pin
narrow SOIC package.
________________________Applications
Local DC-DC Converters for CPUs
Workstations
Desktop Computers
LAN Servers
GTL Bus Termination
_______________Ordering Information
PART
MAX1639ESE
TEMP. RANGE
-40°C to +85°C
PIN-PACKAGE
16 Narrow SO
Pin Configuration appears at end of data sheet.
____________________________Features
o Better than ±1% Output Accuracy Over
Line and Load
o Greater than 90% Efficiency Using N-Channel
MOSFETs
o Pin-Selected High Switching Frequency:
300kHz, 600kHz, or 1MHz
o Over 35A Output Current
o Resistor-Divider Adjustable Output from
1.1V to 4.5V
o Current-Mode Control for Fast Transient
Response and Cycle-by-Cycle Current-Limit
Protection
o Short-Circuit Protection with Foldback Current
Limiting
o Crowbar Overvoltage Protection
o Power-Good (PWROK) Output
o Digital Soft-Start
o High-Current (2A) Drive Outputs
__________ Typical Operating Circuit
VCC
AGND
VDD
BST
TO VDD MAX1639 DH
PWROK
LX
DL
PGND
INPUT
+5V
OUTPUT
1.1V TO 4.5V
REF
FREQ
CC1
CC2
CSH
CSL
FB
________________________________________________________________ Maxim Integrated Products 1
For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800.
For small orders, phone 408-737-7600 ext. 3468.
1 page  
High-Speed Step-Down Controller with
Synchronous Rectification for CPU Power
______________________________________________________________Pin Description
PIN NAME
FUNCTION
1
BST
Boost-Capacitor Bypass for High-Side MOSFET Gate Drive. Connect a 0.1µF capacitor and low-leak-
age Schottky diode as a bootstrapped charge-pump circuit to derive a 5V gate drive from VDD for DH.
2
PWROK
Open-Drain Logic Output. PWROK is high when the voltage on FB is within +8% and -6% of its set-
point.
3
CSL
Current-Sense Amplifier’s Inverting Input. Place the current-sense resistor very close to the controller IC,
and use a Kelvin connection.
4
CSH
Current-Sense Amplifier’s Noninverting Input
5 VCC Analog Supply Input, 5V. Use an RC filter network, as shown in Figure 1.
6
REF
Reference Output, 3.5V. Bypass REF to AGND with 0.1µF (min). Sources up to 100µA for external
loads. Force REF below 2V to turn off the controller.
7 AGND Analog Ground
8 FB Voltage-Feedback Input. The voltage at this input is regulated to 1.100V.
9
CC1
Fast-Loop Compensation Capacitor Input. Connect a ceramic capacitor and resistor in series from
CC1 to AGND. See the section Compensating the Feedback Loop.
10
CC2
Slow-Loop Compensation Capacitor Input. Connect a ceramic capacitor from CC2 to AGND. See the
section Compensating the Feedback Loop.
Frequency-Select Input. FREQ = VCC: 1MHz
11 FREQ
FREQ = REF: 600kHz
FREQ = AGND: 300kHz
12
VDD
Power Input for MOSFET Drivers, 5V. Bypass VDD to PGND within 0.2 in. (5mm) of the VDD pin using a
0.1µF capacitor and 4.7µF capacitor connected in parallel.
13
DL
Low-Side Synchronous Rectifier Gate-Drive Output. DL swings between PGND and VDD. See the
section BST High-Side Gate-Driver Supply and MOSFET Drivers.
14 PGND Power Ground
15 LX Switching Node. Connect LX to the high-side MOSFET source and inductor.
High-Side Main MOSFET Switch Gate-Drive Output. DH is a floating driver output that swings from LX
16 DH to BST, riding on the LX switching-node voltage. See the section BST High-Side Gate-Driver Supply
and MOSFET Drivers.
_______________________________________________________________________________________ 5
5 Page 
High-Speed Step-Down Controller with
Synchronous Rectification for CPU Power
( )VOUT VIN(MAX) − VOUT
L=
VIN(MAX) x fOSC x IOUT x LIR
where f is the switching frequency, between 300kHz
and 1MHz; IOUT is the maximum DC load current; and
LIR is the ratio of AC to DC inductor current (typically
0.3). The exact inductor value is not critical and can be
adjusted to make trade-offs among size, transient
response, cost, and efficiency. Although lower inductor
values minimize size and cost, they also reduce efficien-
cy due to higher peak currents. In general, higher
inductor values increase efficiency, but at some point
resistive losses due to extra turns of wire exceed the
benefit gained from lower AC current levels. Load-
transient response can be adversely affected by
high inductor values, especially at low (VIN - VOUT)
differentials.
The peak inductor current at full load is 1.15 x IOUT if
the previous equation is used; otherwise, the peak cur-
rent can be calculated using the following equation:
( )VOUT VIN(MAX) − VOUT
IPEAK = IOUT + 2fOSC x L x VIN(MAX)
The inductor’s DC resistance is a key parameter for effi-
cient performance, and should be less than the current-
sense resistor value.
Calculating the Current-Sense
Resistor Value
Calculate the current-sense resistor value according to
the worst-case minimum current-limit threshold voltage
(from the Electrical Characteristics) and the peak
inductor current required to service the maximum load.
Use IPEAK from the equation in the section Specifying
the Inductor.
RSENSE =
85mV
IPEAK
The high inductance of standard wire-wound resistors
can degrade performance. Low-inductance resistors,
such as surface-mount power metal-strip resistors, are
preferred. The current-sense resistor’s power rating
should be higher than the following:
IOUT(MAX) 2 x RSENSE
In high-current applications, connect several resistors
in parallel as necessary to obtain the desired resis-
tance and power rating.
Selecting the Output Filter Capacitor
Output filter capacitor values are generally determined
by effective series resistance (ESR) and voltage-rating
requirements, rather than by the actual capacitance
value required for loop stability. Due to the high switch-
ing currents and demanding regulation requirements in
a typical MAX1639 application, use only specialized
low-ESR capacitors intended for switching-
regulator applications, such as AVX TPS, Kemet T510,
Sprague 595D, Sanyo OS-CON, or Sanyo GX series. Do
not use standard aluminum-electrolytic capacitors,
which can cause high output ripple and instability due
to high ESR. The output voltage ripple is usually domi-
nated by the filter capacitor’s ESR, and can be approxi-
mated as IRIPPLE x RESR. To ensure stability, the
capacitor must meet both minimum capacitance and
maximum ESR values as given in the following equa-
tions:
COUT >
VREF 1 +
VOUT
VIN(MIN)
VOUT x RSENSE x fOSC
RESR < RSENSE
Compensating the Feedback Loop
The feedback loop needs proper compensation to pre-
vent excessive output ripple and poor efficiency
caused by instability. Compensation cancels unwanted
poles and zeros in the DC-DC converter’s transfer func-
tion that are due to the power-switching and filter ele-
ments with corresponding zeros and poles in the
feedback network. These compensation zeros and
poles are set by the compensation components CC1,
CC2, and RC1. The objective of compensation is to
ensure stability by ensuring that the DC-DC converter’s
phase shift is less than 180° by a safe margin, at the
frequency where the loop gain falls below unity.
Canceling the Sampling Pole
and Output Filter ESR Zero
Compensate the fast-voltage feedback loop by con-
necting a resistor and a capacitor in series from the
CC1 pin to AGND. The pole from CC1 can be set to
cancel the zero from the filter-capacitor ESR. Thus the
capacitor at CC1 should be as follows:
______________________________________________________________________________________ 11
11 Page | ||
| Páginas | Total 13 Páginas | |
| PDF Descargar | [ Datasheet MAX1639.PDF ] | |
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